The switched mode power supply (SMPS) is a well-known type of power converter having a diverse range of applications by virtue of its small size and weight and high efficiency, for example in personal computers and portable electronic devices such as cell phones. A SMPS achieves these advantages by switching one or more switching elements such as power MOSFETs at a high frequency (usually tens to hundreds of kHz), with the frequency or duty cycle of the switching being adjusted using a feedback signal to convert an input voltage to a desired output voltage. A SMPS may take the form of a rectifier (AC/DC converter), a DC/DC converter, a frequency changer (AC/AC) or an inverter (DC/AC).
There are a number of different circuit topologies for switched mode power supplies, and each has its benefits and shortcomings. Among these are the so-called “fly-back” and “forward” topologies, which are widely used for many different SMPS applications. Fly-back converters are simple, low cost isolated converters that are commonly used for low output power applications. Forward converters, on the other hand, are generally used where greater efficiency and power output are required. The principles of operation of typical fly-back and forward converter circuits will now be explained with reference to FIGS. 1A, 1B, 2A and 2B.
FIG. 1A illustrates a typical fly-back converter with an active clamp on the primary side. More specifically, the fly-back converter 100 comprises a fly-back transformer 110 having a primary winding 110-1 and a secondary side winding 110-2 that are wound around the transformer core 110-3 in a turns ratio N=np/ns, where np and ns are the numbers of turns in the primary winding 110-1 and secondary winding 110-2, respectively. The transformer core 110-3 has a gap (e.g. an air gap) for storing magnetisation energy provided by the primary winding 110-1 during operation.
The primary winding 110-1 is connected to an input voltage source by a primary side circuit comprising a capacitor C1 and actively controlled switches Q1 and Q2. Each of Q1 and Q2 may, as in the present example, be provided in the form of a transistor such as a metal-oxide-semiconductor field-effect transistor (MOSFET) or a bipolar junction transistor (BJT). The secondary side circuit of the fly-back converter 100 comprises a rectifier in the form of diode D2 and a capacitor C2, which are connected as shown in FIG. 1A to rectify and filter the secondary winding voltage.
The fly-back converter 100 can be understood to operate by alternating between two “phases” or “modes” of operation (relating to respective portions of the SMPS switching cycle), as will now be explained.
During the first phase of steady state operation, transistor Q1 is turned ON and transistor Q2 is turned OFF (as shown in traces (a) and (b) in FIG. 1B), so that current from the input voltage source flows through Q1 and the primary winding 110-1 (trace (c)) to magnetise the core 110-3 of the fly-back transformer 110 and store energy in the magnetic field. During the first phase of operation, although current flow through the secondary winding 110-2 is blocked because diode D2 is reverse-biased, current is still supplied to the load of the fly-back converter by the (partial) discharging of capacitor C2, as shown in trace (d) in FIG. 1B.
During the second phase of operation, transistor Q1 is turned OFF and transistor Q2 is turned ON, and the active clamp circuit comprising Q2 and C1 resets the magnetisation of the transformer core 110-3. Meanwhile, on the secondary side, diode D2 becomes forward-biased (since the voltage polarities across the windings have reversed) so that the energy stored in the magnetic field of the transformer 110 during the first phase of operation is released into the secondary side circuit to provide a current to the load and charge the capacitor C2.
As mentioned above, the “forward” topology is also widely used in switching converters, and an example of a forward converter will now be described with reference to FIGS. 2A and 2B. The primary side circuit of the forward converter 200 shown in FIG. 2A is the same as that of the fly-back converter 100 of FIG. 1A, and its description will therefore not be repeated. Furthermore, the transformer 210 of the forward converter 200 is substantially the same as the fly-back transformer 110 but may differ in not having a gap in the transformer core for storing energy. The principal differences between forward converter 200 and the fly-back converter 100 thus lie in the configuration of the secondary side circuit; the secondary side circuit of the forward converter 200 comprises diodes D1 and D2 that are connected to the secondary winding 210-2 and to an inductor (or “choke”) L1 as shown in FIG. 2A, and a capacitor C2. The forward converter 200 can also be understood to operate by alternating between two phases/modes of operation, as follows.
During the first phase of steady state operation, transistor Q1 is turned ON and transistor Q2 is turned OFF (as shown in traces (a) and (b) in FIG. 2B), so that current from the input voltage source flows through Q1 and the primary winding 210-1 (trace (c)) to induce a voltage (scaled by the turns ratio N) and simultaneous current flow in the secondary winding 210-2. The induced voltage causes diode D1 to become forward-biased such that current is allowed to pass from the secondary winding 210-2 to the converter output via the diode D1 and inductor L1 (trace (d) in FIG. 2B), thereby transferring energy from the primary side to the secondary side of the transformer 210. A part of the transferred energy is stored in the magnetic field of inductor L1. Diode D2 remains reverse-biased during the first phase and does not conduct.
During the second phase of operation, when transistor Q1 is turned OFF, current through the filter inductor L1 continues without abrupt change because diode D2, which is forward-biased by the EMF from inductor L1, provides a free-wheeling path for this current (trace (e) in FIG. 2B). Energy stored in inductor L1 during the first phase of operation will thus be released to capacitor C2.
Although converter circuits of the “fly-back” and “forward” topology are widely used, each topology has certain limitations that are hard to resolve.
For example, the fly-back converter does not need a dedicated choke to smooth the output current. Another advantage of a fly-back converter is that it can boost the voltage; if the duty cycle is greater that 50%, the output will be higher than the turns ratio N of the transformer allows. Normally, a fly-back converter operates with a duty cycle just below 50% for maximum efficiency. Operating with a duty cycle above 50% also requires slope-compensation for stability. In other words, in many applications, the fly-back circuit has to be designed outside the “sweet spot”.
Furthermore, although the fly-back converter 100 does not require an output choke, it has the drawback of requiring a relatively large transformer core. In fly-back converters, the gapped transformer inductance results in a zero in the right-half-plane (RHP), which often makes closed-loop compensation in continuous conduction mode (CCM) difficult. Typically, the closed-loop bandwidth in CCM is very narrow and the resulting transient response is very slow. Another drawback of fly-back converters is the requirement of a large output capacitor due to the lack of a second-order low-pass inductor/capacitor filter at the output.
In the forward converter 200, since primary side transistor Q1 can be ON for up to nearly 100% of the time, utilisation of switch components is better than in the fly-back circuit 100 of FIG. 1A; at minimum input voltage (e.g. 36 V), the forward converter 200 operates at almost 100% duty cycle and the primary side transistor Q1 is fully utilised. However, this comes at the expense of needing to provide the forward converter with a choke L1, although the requirements for the output capacitor C2 are reduced. In addition, to avoid transformer saturation, the forward converter 200 requires a magnetisation resetting circuit. All of these increase the component count and manufacturing cost.
In some applications, one of the two topologies described above will clearly be the best choice, while in other applications there is little to choose between them. However, there are also applications where totally different operating conditions are encountered. For instance, the converter may be required to operate in 24 V systems as well as in 48 V systems. Under such circumstances, a converter topology capable of handling a wider input voltage range would be required.
This requirement can be met by the so-called “forward-flyback” (or “fly-forward”) topology, which may be more attractive for handling a wider input voltage range (e.g. 18-72 V) than is normally required (e.g. 36-75 V). A fly-forward circuit topology is employed in Ericsson's PKU 5000E series of DC/DC converters, and will now be described with reference to FIGS. 3A and 3B.
FIG. 3A shows an example of a fly-forward converter 300, which has a primary side circuit that is the same as in the fly-back converter 100 and forward converter 200. However, the transformer 310 of the fly-forward converter 300 has, in addition to a gapped core 310-3, two secondary side windings comprising a first winding 310-2a and a second winding 310-2b, each having the same number of turns as the secondary side winding of the transformers 110 and 210. The secondary side of transformer 310 thus effectively has a larger, centre-tapped secondary winding, which enables the fly-forward converter 300 to swap seamlessly between fly-back to forward modes of operation or to co-operate between them.
More specifically, the fly-forward converter 300 has an output choke L1 connecting the centre-tap terminal of the transformer 310 to the converter's output terminal +OUT. The remaining terminals of the secondary coil are each connected to output terminal −OUT by transistors Q3 and Q4 (although diodes with appropriately arranged polarities could alternatively be used instead of these transistors). The fly-forward converter 300 thus uses more windings on the secondary side than the fly-back and forward converter types since its two transformer windings are connected in series. Transistors Q3 and Q4 must withstand double the voltage that is applied to inductor L1.
During the first phase of steady state operation, transistor Q1 is turned ON and transistor Q2 is turned OFF (as shown in traces (a) and (b) in FIG. 3B) so that current from the input voltage source flows through Q1 and the primary winding 310-1 (trace (c)) to induce a voltage (scaled by the turns ratio N) and simultaneous current flow through transistor Q3 and secondary winding 310-2a. Meanwhile, the core 310-3 is magnetised and stores in the magnetic field some of the energy transferred from the primary side. The voltage induced across winding 310-2a causes current flow from the secondary winding 310-2a to the converter output via the inductor L1 (trace (d) in FIG. 3B), thereby transferring energy from the primary side to the secondary side of the transformer 310. Part of the transferred energy is stored in the magnetic field of L1.
During the second phase of operation, transistor Q1 is turned OFF and transistor Q2 is turned ON, and the active clamp circuit resets the magnetisation of the transformer core 310-3. Meanwhile, on the secondary side, transistor Q4 is turned ON so that the fly-back energy stored in the magnetic field of the transformer 310 during the first phase of operation is released into the secondary side circuit to provide a current to the converter's load and charge capacitor C2.
The fly-forward converter 300 has several advantages over the fly-back and forward converters shown in FIGS. 1 and 2. For example, it can operate with a smaller output capacitance than the fly-back converter 100, and requires no output choke (although an output choke may optionally be included, as in the example of FIG. 3A). Furthermore, there is no RHP zero, so that the converter is easy to compensate and has a fast transient response. In addition, no resetting circuitry is needed, although it may nevertheless be provided, as in the example of FIG. 3A.
These advantages make the fly-forward topology an attractive design choice, particularly for switched mode power supplies that are required to operate over a relatively wide input voltage range.